Upgrade to high rate physical layer by IEEE802.11
Keywords:ieee802.11? cck? ds? direct sequence? frequency hop?
/ARTICLES/1999JUN/1999JUN29_NTEK_RFD_TAC.PDF |
51IIC-Taipei ? Conference Proceedings
Achieving Ethernet
Rates in Wireless LANs
Carl Andren
Senior Principal Engineer
Harris Corporation
Introduction
Just shortly after the IEEE 802.11 standards board approved a
1 and 2Mbps standard for wireless local area networks (WLANs)
in 1997, a working group started working on a higher rate ex-
tension to the physical layer of the standard with the intention
of delivering Ethernet like speeds over existing 802.11 WLAN
systems. This effort was directed at the 2.4 GHz ISM band
which is available almost worldwide and offers 83.5 MHz of
spectrum into which up to 3 channels can be implemented (Foil
2). After months of evaluating various modulation proposals
such as M-ary Orthogonal Keying (MOK), Pulse Position Modu-
lation (PPM), Orthogonal Frequency Division Multiplex
(OFDM), Packet Binary Convolutional Coding, and Orthogo-
nal Code Division Multiplex (OCDM), the working group could
not come to consensus on a single modulation method (Foil 3).
Harris Semiconductor and Lucent Technologies then joined
forces and developed a compromise approach based on Comple-
mentary Code Keying (CCK). In July 1998 the 802.11 work-
ing group adopted CCK as the basis for the high rate physical
layer extension to deliver data rates of 11Mbps. This higher
rate extension was adopted because it easily provides a path for
interoperability with the existing 1 and 2Mbps networks by
maintaining the same bandwidth and incorporating at same pre-
amble and header, which already has a rate shift mechanism.
IEEE 802.11 is not the only group attempting to set standards
for wireless LANs. There are other standards efforts like:
Bluetooth, Home RF Working Group and Personal Area Net-
works that seek to define WLANs for various activities, but
802.11 is the only thrust addressing high data rates for building
wide networks (Foil 4).
Complementary code keying
CCKisavariationofM-aryOrthogonalKeyingmodulationwhich
usesanI/Qdemodulationarchitecturewithcomplexsymbolstruc-
tures. CCK allows for multi-channel operation in the 2.4GHz
ISM band by virtue of using the existing 802.11 1 and 2Mbps
direct sequence spread spectrum (DSSS) channelization scheme.
Thespreadingemploysthesamechippingrateandspectrumshape
as the 802.11 Barker word spread functions allowing for three
(3) non-interfering channels in the 2.4 to 2.483 GHz ISM band.
CCK is an M-ary Orthogonal Keying modulation where one
of M unique (nearly orthogonal) signal codewords is chosen
for transmission (foil 5). The spread function for CCK is cho-
sen from a set of M nearly orthogonal vectors by the data word.
CCK uses one vector from a set of 64 complex (QPSK) vectors
for the symbol and thereby modulates 6-bits (one-of-64) on each
8 chip spreading code symbol. Two (2) bits are sent by QPSK
modulating the whole code symbol and this modulates allows
for 8-bits onto each symbol. The formula that defines the CCK
codewords is shown in Foil 6. In it, there are 4 phase terms.
One of them modulates all of the chips and this is used for the
QPSK rotation of the whole code vector. The others modulate
every odd chip, every odd pair of chips and every odd quad of
chips respectively.
Walsh functions were used for the M-ary Bi-Orthogonal key-
ing (MBOK) modulation first proposed by Harris. They are
the most well known orthogonal BPSK vector set and available
in 8 chip (powers of 2) vectors. To transmit enough bits per
symbol, the MBOK modulation was used independently on the
I and Q channels of the waveform effectively doubling the data
rate. CCK on-the-other-hand uses a complex set of Walsh/
Hadamard functions known as Complementary Codes. Walsh/
Hadamard properties are similar toWalsh functions but are com-
plex, that is, more than two phase, while still being nearly or-
thogonal. With complex code symbols, we cannot transmit si-
multaneous independent code symbols. Since the set of comple-
mentary codes is more extensive, however, we have a larger set
of nearly orthogonal codes to pick from and can get the same
number of bits transmitted per symbol. Additionally, the
multipath performance of CCK is better than MBOK due to the
lack of cross rail interference as will be explained later.
For MBOK, there are 8 BPSK chips that have a maximum
vector space of 256 code words of which you can find sets of 8
that are orthogonal. Two independent BPSK vector sets are se-
lected for the orthogonal I and Q channels which modulate 3-
bits on each. Two additional bits are used to BPSK modulate
each of the spreading code vectors. For CCK, there are 65536
possible code words, and sets of 64 that are nearly orthogonal.
This is because it really takes 16 bits to define each code vector
(Foil 5). To get the 5.5 Mbps data rate, a subset of 4 of the 64
vectors that have superior coding distance is used.
One of the advantages of CCK over MBOK is that it suffers
less from multipath distortion in the form of cross coupling of I
and Q channel information. The information in CCK is en-
coded directly onto complex chips which cannot be cross-couple
corrupted by multipath since each channel finger has an Aej
distortion. A single channel path gain-scales and phase-rotates
the signal. A gain scale and phase rotation of a complex chip
still maintains I/Q orthogonality. This superior encoding tech-
nique avoids the MBOK, corruption resulting from encoding
Mark Webster
Staff Engineer
Harris Corporation
52 IIC-Taipei ? Conference Proceedings
half the information on the I-channel and the other half on the
Q-channel, which easily cross-couple corrupts with the
multipath's Aej
phase rotation.
In an overview (Foil 7) of the CCK modulation modes and
the original 1 and 2 Mbps modes which are used in the Harris
HFA3860B chip. For 1 Mbps, the signal is modulated BPSK
by one bit per symbol and then spread by BPSK modulating
with the 11 Mchip/s Barker code. For 2 Mbps, the signal is
QPSK modulated by two bits per symbol and then BPSK spread
as before. For the 5.5 Mbps CCK mode, the incoming data is
grouped into 4 bit nibbles where 2 of those bits select the spread-
ing function out of the set of 4 while the remaining 2 bits QPSK
modulate the symbol. The spreading sequence then DQPSK
modulates the carrier by driving the I and Q modulators. To
make 11 Mbps CCK modulation, the input data is grouped into
2 bits and 6 bits. The 6 bits are used to select one of 64 com-
plex vectors of 8 chip length for the symbol and the other 2 bits
DQPSK modulate the entire symbol.
802.11 interoperability
Interoperability was a priority amongst the 802.11 working
group in the selection of the waveform for higher rates. In par-
ticular, the signal acquisition scheme for 802.11 uses a specific
preamble and header using the 1 Mbps modulation
and has provision built-in for sending the payload at
higher rates. The packet frame structure and protocol
of 802.11 much like 802.3 Ethernet, however it must
operate wirelessly in a harsh RF environment. This
means that the signal levels may become corrupted
and subject to multipath. Signal acquisition and syn-
chronization of the preamble and header are critical.
The preamble and header consists of five (5) fields.
They are: Preamble, SFD, Signal (rate), Service,
Length and CRC. The header takes 48 bits, and the total length
of the acquisition sequence is 192 microseconds. The preamble
and header is modulated using the 1 Mbps modulation rate and
is scrambled with a self synchronizing scrambler. The high
rate scheme will initially use this acquisition sequence which
already has a rate field that can be programmed for 1, 2, 5.5 or
11 Mbps. The high rate 802.11 standard is being written with
an optional shorter acquisition sequence for lower packet over-
head (Foil 8).
The 802.11 packet transmission protocol is Carrier Sense
Multiple Access with Collision Avoidance (CSMA/CA). This
differs from "wired" Ethernet which uses collision detection.
Radios can't detect collisions, therefore they use collision avoid-
ance using a listen before talk and random back off deferral
mechanisms. Since all stations use the same acquisition se-
quence at the lowest basic rate, all stations can see the traffic
and process the signals at the appropriate rate. If legacy 1 and
2Mbps stations receive the packet header, but are not capable
of processing the higher rate, they can still defer the medium
based on knowing that an 802.11 signal has been sensed and
knowing the length of time it is on the air.
To insure that the modulation has the same bandwidth as
the existing 802.11 DS modulation, the chipping rate is kept at
11 Mchip/s while the symbol rate is increased to 1.375 MSps.
This accounts for the shorter symbols and makes the overall bit
rate 11 Mbps. This approach makes system interoperability
with the 802.11 preamble and header much easier. The spread
rate remains constant and only the data rate changes and the
spectrum of the CCK waveform is same as the legacy 802.11
waveform.
Walsh and complementary codes
Walsh functions have a regular structure and at least one mem-
ber that has a substantial DC bias. In this case it is the first row
with all 1s. All the rest are half 1s and half 0s. The DC bias can
be reduced on the worst member of the set by multiplying all
members with a cover code. This, however introduces a smaller
DC bias in half of the members of the set.
The main problem of MBOK is caused by the fact that it uses
independent codes on the in-phase and quadrature signals, which
creates a significant amount of cross-rail interference in the pres-
ence of multipath. To avoid this, one would ideally transmit
only symbols for which processing could be done on I and Q
simultaneously, and use code words that all have good
autocorrelation properties, such that there is minimal inter-sym-
bol and inter-chip interference. Such codes actually exist in the
form of the complementary codes. For a code length of 8 chips,
256 possible sequences c can be constructed as follows, using
4 QPSK phases 1 to 4.
Complementary Codes
Note that 1 is present in all 8 chips, so it simply rotates the
entire code word. Hence, to decode these code set, one would
need 64 correlators plus an additional phase estimation of the
code that gave the largest correlation output. The correlation
can be significantly simplified by using techniques like the fast
Walsh transform (analogous to an FFT butterfly circuit). In
fact, when the 4 input phases 1 to 4 are binary, then the
complementary code set reduces to a modified Walsh code set,
similar to the one used in Harris' original proposal.
CCK codes have a number of advantages. They are: 1)They
are nearly orthogonal, yielding superior error-rate performance.
2) Using a cover code eliminates one bad DC bias member,
without destroying orthogonality.
Implementation
Harris HFA3860B implements the legacy 802.11 1 and 2Mbps
DSSS demodulation, MBOK, and the CCK waveform at
11Mbps as adopted by the 802.11 working group This chip is
part of the overall PRISM chip set as shown in foil 9. The
processing of the waveform is carried out in the Baseband Pro-
cessor (Foil 10). The Barker coded signals such as the pre-
amble are correlated in two time invariant matched filter
correlators. This allows rapid acquisition of the preamble and
is also used for demodulation of the BPSK and QPSK modu-
lated 1 and 2 Mbps signals. The demodulator converts the
correlator outputs from the Cartesian coordinate system to the
polar coordinate system. All demodulation processes are there-
fore non complex. For the high rate modes, the signal is de-
rotated by a complex multipliery and then correlated with a
Fast Walsh Transform block (FWT). This is followed by a big-
gest picker and DQPSK demodulator. Carrier tracking in these
modes used decision directed phase detection and a lead-lag
filter in the tracking loop.
Fast transform structure
The four phase variables each take on values of [0, /2, , 3/
2], and there are 256 (4*64) possible 8 chip codes. These codes
have an inherent "Walsh" type structure that allow a simple
53IIC-Taipei ? Conference Proceedings
butterfly implementation of the decoder (Foils 11 and 12). Al-
though it is possible to squeeze a few more complementary codes
out of this 8 chip set, the rest of the codes cannot be decoded
with the modified fast Walsh transform. Foil 11 shows the ba-
sic fast Walsh block which brings in 8 chips of soft decision
data shown here by x0, x1, x 7, and produces 16 possible corre-
lation for given values 1 and 2.
Foil12showsall256possiblecorrelatoroutputs. TheBFWB's
are shown in detail in Foil 11 There are 28 butterflies needed for a
length 8 transform. Each butterfly requires 4 additions (the phase
rotationsaretrivialfor4-PSK),sothetotalnumberofoperationsis
112complexadditions.Thedirectcalculationmethodwith64sepa-
rate correlators requires 512 complex additions, so the fast trans-
form reduces the complexity by almost a factor of 5.
CCK is inherently a quadrature MOK signal. For the full
data rate potential, we QPSK modulate the starting phase of the
symbols to get 11 Mbps as shown in Foil 6. To reduce the data
rate for a more robust lower data rate, we can trim the signal set
to one that has the greatest distance properties with a reduced
number of vectors. For 5.5 Mbps, there are two options --
first, trim the 64-ary set to 8-ary and BPSK modulate the sym-
bols or second, trim the set to 4-ary and QPSK modulate the
symbols. Either scheme achieves 4 bits per symbol but simula-
tions conclude that the latter is more robust in multipath.
Range performance
The CCK modulation achieves excellent range due to the fact
that MOK has better Eb/N0 performance than BPSK. This per-
formance is due the embedded coding properties of the spread-
ing modulation. The modulation basically ties several bits to-
gether so that the receiver makes a symbol decision. If a sym-
bol is in error then all of the bits in that symbol are suspect, but
not all will necessarily be in error. Thus, the symbol error rate
and the bit error rates are related. While the SNR required to
make a symbol decision correctly is higher than required to
make a one bit decision, it is not as high as required to make all
of the bit decisions of a symbol independently and correctly.
Thus, some coding gain is embedded in the basic spreading
waveform. Simulations conclude CCK modulation yields
achievable ranges of 100' reliably and that the high rates are
more susceptible to multipath than the lower rates as would be
expected from the higher required Es/N0.
Performance parameters
The FCC requires that DSSS modulations used in the 2.4 GHz
ISM band have a minimum processing gain of 10 dB. The
spreading/despreading operations using CKK does provide 11
dB of processing gain when used in accordance with the FCC
rules. The reduction in bandwidth provides 9 dB and MOK
coding constitutes 2 dB of coding gain. After de-spreading, the
SNR improves by 11 dB over the SNR in the spread bandwidth.
Under these conditions radios designed with CCK modulation
satisfies the FCC's requirements with ample margin for the CW
jamming test.
Antenna diversity helps insure a reliable 11 Mbps link for
indoor environments. Any high rate modulation is more sus-
ceptible to multipath interference and filter distortion than lower
rate modulations due to the higher required SNR (Es/N0). Ex-
tensive testing concluded that CCK modulation in the indoor
environment and have shown acceptable performance. How-
ever using antenna diversity yields significant improvement in
the packet error rate.
For high multipath environments, such as factory and manu-
facturing plants, a CCK demodulator using a RAKE receiver
can tolerate delay spreads of 100nsec and >100nsec when com-
bined with a decision feedback equalizer (DFE).
Multipath
Stressed links can be substantially improved by equalization
where substantial multipath is encountered. The typical envi-
ronment for wireless LANs is the office or home. There, the
delay spread is on the order of 100 ns or less as shown in Foil
13. Usually, the presence of walls in the direct path makes the
system work from indirect paths and that makes the impulse
response have energy leading the peak of the energy. This is
called precursor energy and requires more complex processing
that does the trailing energy from delayed echoes. Typically,
precursor processing involves complex multiplies whereas, trail-
ing energy involves adds and subtracts.
Large warehouses and factories (Foil 14) often have much
larger delay spreads and this takes more equalization process-
ing. There is a range of complexities in the receive processing
that can be employed to meet each of these environments. While
it is desirable to make one chip that can handle all cases, it is
often impossible to meet cost and power goals if too much ca-
pability is taken on.
RAKE receiver
The RAKE receiver principle is good for modest multipath of
around 100 ns delay spread. Foil 15 shows how the RAKE
receiver is implemented in the PRISM chip under development.
The classical RAKE receiver has multiple correlators with a
delay and combine circuit following the correlators.
For the CCK waveform, this would result in an unduly com-
plex design as the CCK scheme requires multiple correlators
for each of the multiple correlators of the RAKE technique. By
transformation, the RAKE combiner can be moved to the input
of the correlator bank where it is much simpler. In this form, it
is called a Channel Matched Filter, and it complements the chan-
nel impulse response and therefore corrects for it. This removes
the channel effects as far as can be done with a fixed filter, but
does not correct for inter symbol or inter chip interference (ISI/
ICI). Simulations have shown that the RAKE only receiver can
achieve near 100 ns delay spread performance without an equal-
izer.
RAKE plus ISI receiver
For the larger delay spreads of the factory environment, an ISI/
ICI equalizer is needed and that raises the complexity in sev-
eral ways. First, the equalizer requires lots of gates running
very fast in the receiver, and second it needs decision feedback
to properly handle the ISI and ICI.
The first stage of equalization is ISI cancellation and that
involves taking the output of the symbol decisions and then
subtracting the left over energy from the previous symbol from
the current symbol before demodulation the current one. As
shown in Foil 16, the past decisions are weighted with the chan-
nel impulse response and subtracted from the input signal to
the CMF.
RAKE plus ISI/ICI receiver
The next step in equalization is canceling the ICI interference
and that takes a more complex process since the ICI depends
on which of the 64 vectors was received. The Harris technique
embeds the ICI equalizer into the FWT correlator block since
the correction does not need to be performed on the fly, but can
be set up prior to reception once the CIR is known. The full
receiver with RAKE and ISI/ICI equalizer is shown in Foil 17.
It has performance in multipath out to 333 ns for the most de-
manding applications.
54 IIC-Taipei ? Conference Proceedings
Equalizer performance
The performance of each of the options shown above has been
simulated (Foil 18) and shows that the receiver can be tailored
in complexity for meeting all the various environments that a
WLAN is likely to encounter.
Future developments
Harris has embarked on the design of new chip sets to reduce
the cost of WLANS while increasing their capability. The
PRISM 98 suite is designed with greater levels if integration
while also adding the RAKE and equalizer functions in a staged
development. (foils 19 to 21).
Authors' contact details
Carl Andren and Mark Webster
Harris Semiconductor
2401 Palm Bay Road, N.E.
MS:62A-024
Palm Bay, Florida
Tel: 407-724-7535
email: candren@harris.com/mwebster@harris.com
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